The invention relates to a drive controller for controlling brushless motors in use as motors for driving discs of audio systems, video tape recorders, and personal computers.
This type of controllers for brushless motors is described, for example, in Japanese Patent Laid-Open Publication No. 10-146085. This conventional drive controller has a structure as shown in FIG. 1. The controller is operated using various control signals as shown in FIG. 2 controlling various components thereof as will be discussed in more detail below.
In the prior art, a brushless motor M has a permanent magnetic rotor and three-phase armature coils U, V, and W arranged on a circumference of a stator. The stator has rotor-position detectors, one for each of the armature coils, located at the respective armature coils. For brevity, these rotor-position detectors 11 are shown in FIG. 1 altogether outside the brushless motor M.
As shown in FIG. 1, transistor switches for the three-phase motor M consist of P-channel type Metal Oxide Semiconductor field effect (MOS) transistors QUH, QVH, QWH connected to the positive potential source VDD (such transistors hereinafter referred to as positive side transistors), and N-channel type MOS transistors QUL, QVL, QWL connected to the ground (such transistors hereinafter referred to as negative side transistors). The P-channel type and N-channel type transistors are turned ON/OFF by the gate controlling signals supplied to the respective gates of the transistors.
The rotor-position detector 11 may be formed of, for example, hall elements to generates three xe2x80x9cpositive phasexe2x80x9d sinusoidal signals in response to the rotating magnetic field created in the U, V, and W coils, and three sinusoidal xe2x80x9cnegative phasexe2x80x9d signals which are the inversions of the respective positive phase signals. The three phase signals are out of phase with each other by 120xc2x0 (=360xc2x0/3 coils).
A position-detection/phase-shifting circuit 14 takes a difference between the positive and negative outputs contained in the respective three-phase outputs HU, HV, and HW received from the rotor-position detector 11 to eliminate noises that exist in common in the signal lines. Thus, the position-detection/phase-shifting circuit 14 generates six difference signals HU-HV, HV-HW, HW-HU, HV-HU, HW-HV, and HU-HW. From these signals, phase-shifted signals HU1, HV1, and HW1 having a phase difference xcex94xcex8=30xc2x0, for example, can be generated as shown in FIGS. 2(a)-(c). These phase-shifted signals HU1, HV1, and HW1 serve as position detection signals. Each of the phase-shifted signals HU1, HV1, and HW1 is compared with the inverted version of the respective phase-shifted signals to generate polarity-determination signals UHL, VHL, and WHL for the respective U, V, and W phases, respectively.
The reason for forming three phase-shifted signals HU1, HV1, and HW1 is as follows. There is a delay between the point when the armature of the motor M is actuated by a signal received from the rotor-position detector 11 and the point when the armature is actually energized by an energizing current that flows therethrough, due to the inductance of the armature. The delay depends on the time constant, which is determined by the inductance of the armature. Thus, the commutation time of the current through the armature delays behind the normal commutation time, which can harm the driving efficiency of the motor and entail fluctuation of torque.
A full-wave rectifier 15 rectifies the phase-shifted signals HU1, HV1, and HW1 received from the position-detection/phase-shifting circuit 14 and generates three full-wave outputs HU2 (FIG. 2(d)), HV2, and HW2, which are fed to a comparator 16. The reference potential of the HU2, HV2, and HW2 is taken to be the ground potential Vgnd. Waveforms of the signals HV2 and HW2 are not shown in FIG. 2.
The oscillator 13 includes a built-in triangular signal generation circuit for generating high frequency triangular signals OSC (FIG. 2(e)) in the frequency range above 16 KHz for example, which are fed to the comparator 16. The triangular signal generation circuit includes an operational amplifier, a constant current power supply, and a condenser. The triangular signals OSC also has a reference voltage equal to the ground potential Vgnd.
The comparator 16 receives rectified full-wave signals HU2, HV2, HW2 and triangular signals OSC from the oscillator 13, and compares them to generate pulse width modulation (PWM) signals UPWM, VPWM, and WPWM from their differences.
Pre-drive circuits 17U, 17V, and 17W provided for the respective three phases receive PWM signals UPWM, VPWM, WPWM from the comparator 16, and polarity discrimination signals UHL, VHL, and WHL for the respective phases from the position-detection/phase-shifting circuit 14 directly or via the comparator 16. These pre-drive circuits invert or switches PWM signals UPWM, VPWM, WPWM in accordance with the polarity discrimination signals UHL, VHL, WHL to form gate control signals VUGH, VUGL, VVGH, VVGL, VWGH, and VWGL as shown in FIGS. 2(f)-(k), which control signals are supplied to the P-channel type MOS transistors QUH, QVH, and QWH on the positive side, and to the N-channel type MOS transistors QUL, QVL, QWL on the negative side.
As a typical example, take gate control signals VUGH and VUGL for the U-phase. It is seen in FIG. 2 that in the first half period shown of FIG. 2, the positive side MOS transistor QUH is turned ON and OFF by the gate control signal VUGH, while the negative side MOS transistor QUL is turned OFF by the gate control signal VUGL. In the second half period, on the other hand, the positive side MOS transistor QUH is turned OFF by the gate control signal VUGH, while the negative side MOS transistor QUL is turned ON and OFF by the gate control signal VGUL. It is noted that in both the positive side MOS transistor QUH and the negative side MOS transistor QUL, a backward current flows through a back gate contact of the transistor and a parasitic diode associated with it during the OFF periods.
A torque instruction circuit 12 generates control signals for controlling the rotational speed of the motor M by controlling the amplitudes of the phase-shifted signals HU1, HV1, and HW1. This can be done by comparing a measured value Vdet indicative of the actual speed of the motor M and a preset value Vs and controls the amplitudes in accordance with the differences between them.
In this arrangement, if, for example, the measured value Vdet representing the actual speed of the motor M is greater than the preset value Vs (i.e. the motor is faster than the intended speed), the torque instruction circuit 12 supplies the position-detection/phase-shifting circuit 14 with a control signal determined by the difference between them to reduce the amplitudes of the phase-shift signals HU1, HV1, HW1. Accordingly, the rectified full-wave signals HU2, HV2, HW2 are output from the full-wave rectifier 15 with a reduced wave height.
The reduction of the height of the rectified full-wave signals HU2, HV2, HW2 results in a decrease in the pulse width of the PWM ON-OFF duty pulses UPWM, VPWM, WPWM issued from the comparator 16, which in turn decreases the current energizing the motor M via the MOS transistors QUH QWL for the respective U, V, and W phases. If the rotational speed is lower than the preset value, the current to the motor is increased by the same mechanism to thereby raise the speed of the motor. In this manner, the motor speed is controlled precisely.
In this conventional drive controller, the gate control signals VUGH VWGL shown in FIGS. 2(f)-(k) are supplied to the MOS transistors QUH QWL in the respective phases, such that the negative side MOS transistors are turned OFF while the positive side MOS transistors are turned ON and OFF while PWM controlled in the respective phases. Conversely, the positive side MOS transistors are turned OFF while the negative side MOS transistors are turned ON and OFF during the PWM control.
FIGS. 3 and 4 illustrate PWM control in the U-phase when the positive side MOS transistor QUH is turned ON and OFF in the PWM control, while the negative side MOS transistor QUL is turned OFF. It is noted that, in this instance, a drive current I flows in the direction shown by a solid arrow when the positive side MOS transistor QUH is ON, and a regeneration current I flows in the direction as indicated by a broken arrow when the positive side MOS transistor is OFF, as shown in FIG. 3. Since the negative side MOS transistor QUL is turned OFF, the regeneration current I flows through a parasitic diode Pd via the back gate contact of the negative side MOS transistor QUL. A similar regeneration current will take place when the conduction status of the positive side and negative side MOS transistors are reversed, and the same is true with other MOS transistors in V and W-phases.
This implies that the power loss due to the regeneration current I through the parasitic diode Pd is fairly large since the potential drop Vf by the parasitic diode Pd is about 0.7 Volts. Unfortunately, this takes place continuously during the PWM control.
In conventional drive controller, rectified full-wave signals HU2, HV2, HW2 received from a full-wave rectifier 15 are compared with the oscillation signals OSC received from the oscillator 13 to form PWM signals UPWM, VPWM, WPWM in the comparator 16. It is noted that, in these comparison, the reference potential is set to the ground potential Vgnd.
In particular, the oscillator 13 generates triangular signals, by using the operational amplifier, the constant current power supply, and the condenser, as a basis for the switching operation to increase/decrease the electric current, wherein the switching is performed at an upper and a lower peaks (upper and lower apexes) of a triangular wave. In this case, however, generation of triangular waves cannot be accurately controlled since the magnitude of overshooting varies with the frequency of the oscillator 13 and since different characteristics of the constant current power supplies on the positive and the negative sides as well as variations in the manufacturing processes all affects the triangular signals generated.
Under these conditions, the triangular signals OSC of the oscillator 13 can have a blind zone above the ground potential in case that the minimum voltage (of the lower edges or lower apexes) of the triangular signals does not reach the ground potential Vgnd, as shown in FIG. 5, or a portion of the lower edges remain at the ground potential Vgnd, thereby deforming the triangular signals, as shown in FIG. 6.
If such a blind zone takes place or the lower edges of the triangular signals stays at the ground potential, the oscillator cannot provide sufficient output power even under a torque instruction, or minimum pulse width of the output signals may become very large. If this happens, the motor cannot be properly controlled.
It is therefore an object of the invention to provide a drive controller for brushless motors capable of performing PWM control with a reduced power loss caused by regeneration current during the PWM control, without inducing any blind zone or distortion in the output of the PWM control oscillator.
In accordance with one embodiment of the invention, there is provided a drive controller for brushless motors, comprising:
a multiplicity of switching means each having
a first switching element connected between a first power supply providing a first potential and an output terminal of the drive controller, and
a second switching element connected between the output terminal and a second power supply providing a second potential;
oscillator means for generating triangular signals;
position signal generation means for generating sinusoidal position detection signals based on the signal detected by a position detector of the brushless motor; and
comparator means for comparing the magnitudes of the triangular signals with the sinusoidal position detection signals to generate PWM signals, wherein the first switching element and the second switching element of each switching means are turned ON/OFF alternately by said PWM signals.
The first and the second switching elements of each switching means can be a P-channel type MOS transistor and an N-channel type MOS transistor, respectively.
Each of the PWM signals generated by the comparator means may be split to two signals, one for each of the first and the second switching elements when the first and the second switching elements are P-channel type MOS transistor and N-channel type MOS transistor, respectively.
In this drive controller for a brushless motor, when one of these switching elements, for example the first one, QUH, is turned ON and OFF by a PWM signal, the other one, i.e. the second switching element QUL, will be conversely turned OFF and ON. That is, in the PWM control, the second switching element QUL will be turned OFF when the first switching element QUH is turned ON, and vice versa. Consequently, a regeneration current of the motor will flow through the conducting switching element, not through a parasitic diode Pd as in the conventional switches. The voltage drop Von across the conducting transistor due to its resistance (said resistance hereinafter referred to as ON-resistance) is in the range of 0.1-0.2 Volts, which is much smaller than the voltage drop Vf (about 0.7 Volt) of a parasitic diode Pd. Hence, the power loss due to the regeneration current can be reduced accordingly.
It is noted that, in the arrangement described above, sinusoidal position detection signals from the position signal generation means 14 are used in the comparator means as they are (i.e., without being rectified) to generate a PWM signal. The PWM signals may be applied to the first and the second switching elements (e.g. QUH and QUL, respectively) without inverting them in phase, or applied with their phases inverted, depending on the channel type of the switching elements used. In this instance, polarity discriminating signal is not needed, as is needed in conventional controllers.
In the invention, the mean or central potential of the triangular signals generated by the oscillator means can be set to the same mean potential of the sinusoidal position detection signal generated by the position signal generation means. The controller may be provided with a constant voltage generation means that generates a first potential, a second potential which is higher than the first potential by a predetermined potential, a third potential which is lower than the first potential by a predetermined potential so that the mean potential of the sinusoidal position detection signal and the mean potential of the triangular signals are set to the first potential by feeding the first potential to the position signal generation means, and by feeding the second and the third potentials to the oscillator means. Thus, the oscillator means has the second potential as its upper limit and the third potential as its lower limit.
In this arrangement, the triangular signals do not involve a blind zone or a distortion, since the triangular signals are not offset above the ground potential Vgnd nor stays at the ground potential as in the conventional controller. It is noted that the mean potential of the triangular signals and the sinusoidal position detection signals are set to coincide. Thus, the output duty of the PWM signals can be fully utilized over the range between 0 and 100%, so that fine drive control can be effected, even for a minute rotation.